(a) Field of the Invention
The present invention relates to a quadrature modulator and, more particularly to a quadrature modulator which performs modulation of quadrature (orthogonal) carrier waves with a digital baseband signal to deliver an output digital carrier signal. The present invention also relates to a method for quadrature-modulating carrier waves with a digital baseband signal.
(b) Description of the Related Art
In a wireless transmitter block of a digital cellular phone, for example, orthogonal carrier waves each having a frequency equal to the frequency of the output digital carrier signal are quadrature-modulated with a digital baseband signal which includes information to be transmitted. The modulated carrier waves are then added together to generate the output digital carrier signal, and transmitted through a transmission antenna. This scheme of quadrature modulation is suitable for simplification of the communication system and for reduction of noise in the transmitted carrier signal.
FIG. 1 shows a fundamental structure of a conventional quadrature modulator, wherein the output frequency from a local oscillator 400 is exactly equal to the frequency of the output digital carrier signal delivered from the modulation block 200. This causes affection of the local oscillator 400 by the output carrier signal fed back through the transmission antenna to degrade the modulation accuracy. Thus, a metallic shield is generally provided for encircling the quadrature modulator as a whole.
FIG. 2 shows another quadrature modulator, wherein the above problem is solved by the difference between the oscillation frequencies of a pair of local oscillators 401 and 501 and the frequency of the output carrier signal. However, since the frequency mixer 600 has a non-linearity, a plurality of harmonics (harmonic signals) of the oscillation frequencies of the local oscillators 401 and 501 are generated during the frequency conversion by the frequency mixer 600. The harmonics, also subjected to frequency conversion, generate spurious signals in the vicinity of the output carrier signal.
As a practical example of the frequencies used in mobile stations of personal digital cellular (PDC) system prescribed in the standards of cellular phones in Japan, the output frequencies of the local oscillators 401 and 501 are 135 and 795 MHz, respectively. In this case, the frequency of the output digital carrier signal is 930 MHz. The frequencies of spurious signals occurring in the most vicinity of the frequency 930 MHz of the output carrier signal are 945 MHz and 915 MHz. The frequency 945 MHz is seventh-order harmonic of 135 MHz, and the frequency 915 MHz is a frequency difference between the second-order harmonic of 795 MHz and the fifth-order harmonic of 135 MHz.
These spurious signals occur within or in the vicinity of the band of the output digital carrier signal, and are difficult to remove by using filters, acting as interference waves against the adjacent transmission channels or other communication systems.
FIG. 3 shows another quadrature modulator described in Patent Publication JP-A-10-4437, which solves the above problem in the quadrature modulator of FIG. 2. The quadrature modulator of FIG. 3 includes a local oscillator 402 for oscillating at a specified frequency, a first ½-frequency-divider 310 for dividing the output frequency of the local oscillator 402 by a factor of two, a second ½-frequency-divider 350 cascaded from the first ½-frequency-divider 310 for dividing the output frequency thereof by a factor of two, a frequency mixer 320 for frequency conversion using the output frequencies of second ½-frequency-divider 350 and the local oscillator 402, a band-pass-filter (BPF) 330 for removing the image signal from the output of the frequency mixer 320, a frequency-multiplier (doubler) 250 for doubling the output of the BPF 330, a third ½-frequency-divider 240 for dividing and phase-shifting the output from the frequency multiplier 250 to output a pair of orthogonal carrier waves having a phase difference of 90 degrees therebetween, first and second multipliers 210 and 220 for modulating the carrier waves with a baseband signal generated by a digital signal generator 101, and an adder 230 for adding the outputs of the first and second multipliers 210 and 220 to generate an output digital carrier signal.
In operation, the first frequency divider 310 divides the output oscillation frequency from the local oscillator 402 by a factor of two, and the second frequency divider 350 divides the output of the first frequency divider 310 by a factor of two to deliver its output to the frequency mixer 320. The frequency mixer 320 acts for frequency conversion by using the output frequencies from the local oscillator 402 and the second frequency divider 350.
FIG. 4 shows an example of the frequency mixer, which is implemented by a so-called double-balanced mixer. The input signals are supplied to both the input terminals Vin1 and Vin2, whereas the input terminals Vin1b and Vin2b are grounded through a capacitor, or may be applied with the inverted input signals.
Assuming that the outputs from the local oscillator 402 and the second frequency divider 350 are expressed by VHsinωosct and VLsinωosct/4, respectively, the output LO(t) of the frequency mixer 320 is expressed as follows:
                              LO          ⁡                      (            t            )                          =                ⁢                              V            H                    ⁢          sin          ⁢                                          ⁢                      ω            osc                    ⁢          t          ×                      V            L                    ⁢          sin          ⁢                                          ⁢                      (                                          ω                osc                            ⁢                              t                /                4                                      )                                                  =                ⁢                                                            -                                  (                                      1                    /                    2                                    )                                            ·                              V                L                                      ⁢                          V              H                        ⁢            cos            ⁢                                                  ⁢                          (                                                ω                  osc                                +                                                      ω                    osc                                    /                  4                                            )                        ⁢            t                    +                                                ⁢                                            (                              1                /                2                            )                        ·                          V              L                                ⁢                      V            H                    ⁢          cos          ⁢                                          ⁢                      (                                          ω                osc                            -                                                ω                  osc                                /                4                                      )                    ⁢          t                                        =                ⁢                                                            -                                  (                                      1                    /                    2                                    )                                            ·                              V                L                                      ⁢                          V              H                        ⁢            cos            ⁢                                                  ⁢                          (                              5                ⁢                                                      ω                    ⁢                                                                                                  osc                                ⁢                                  t                  /                  4                                            )                                +                                                ⁢                                                            (                                  1                  /                  2                                )                            ·                              V                L                                      ⁢                          V              H                        ⁢            cos            ⁢                                                  ⁢                          (                              3                ⁢                                  ω                  osc                                ⁢                                  t                  /                  4                                            )                                ,                    wherein the gain of the double-balanced mixer is assumed at “1” for purpose of simplification.
That is, a pair of angular frequency components 5ωosc/4 and 3ωosc/4 are generated therein.
Assuming that the output digital carrier signal has a frequency of 930 MHz, as in the case of the quadrature modulator of FIG. 2, the local oscillator 402 delivers an output frequency of 1240 MHz to the first frequency divider 310. The first frequency divider 310 delivers an output frequency of 620 MHz to the second frequency divider 350, which delivers an output frequency of 310 MHz. The frequency mixer 320 delivers a signal having frequency components of 930 MHz and 1550 MHz based on the output frequency of 1240 MHz from the local oscillator 402 and the output frequency of 310 MHz from the second frequency divider 350. In this case, the difference between the frequency (3ωosc/4: 930 MHz) of the carrier wave and the frequency (5ωosc/4: 1550 MHz) of the image signal is 620 MHz.
The BPF 330 removes the image signal having the frequency component of 1550 MHz, passes the carrier frequency component of 930 MHz. The frequency doubler 250 then doubles the output from the BPF 330 to deliver an output frequency of 1860 MHz. The third frequency divider 240 then divides and shifts in phase the output from the frequency doubler 250 to deliver a pair of carrier waves having a frequency of 930 and a phase difference of 90 degrees therebetween. The first and second multipliers 210 and 220 modulates the carrier waves with the digital baseband signal output from the digital signal generator 101 to output modulated signals, which are added in the adder 230 to be delivered as an output digital carrier signal. The frequency of each block in the quadrature modulator of FIG. 3 is shown in FIG. 5 in terms of the output frequency fosc of the local oscillator 402.
In the quadrature modulator of FIG. 3, the output digital carrier signal has a frequency of 3fosc/4 which is different from the output frequency fosc of the local oscillator. This prevents degradation of the modulation accuracy, which is encountered due to the affection by the feed-back of the output carrier signal through the transmission antenna in the quadrature modulator of FIG. 2.
Spurious signals may be generated in the frequency mixer 320 as harmonics of the signal having a ¼-divided frequency of the output frequency of the local oscillator 402 due to the non-linearity of the frequency mixer 320. However, these spurious signals do not act as interference waves against the output carrier signal because the spurious signal among these spurious signals which has a frequency in the vicinity of the carrier frequency has a frequency equal to the carrier frequency itself.
The quadrature modulator of FIG. 3, however, has the drawback of a complicated structure including a large number of constituent elements therein. For example, the frequency doubler 250 is provided for the third ½-frequency-divider 240 which delivers a pair of orthogonal carrier waves while dividing the input thereof by a factor of two, The frequency doubler 250 implemented by the double-balanced mixer shown in FIG. 4 receives the output from the BPF 330 through two input terminals Vin1 and Vin2. For the double-balanced mixer receiving the same frequency signal through the two inputs, a d.c.-blocking capacitor should be provided for suppression of a d.c. offset voltage which occurs based on the phase difference between the two input terminals. In addition, as shown in FIG. 3, there are three ½-frequency-dividers. The complicated structure increases the dimensions of the IC pellet.
The quadrature modulator of FIG. 3 has another drawback in connection with the BPF 330. Specifically, the BPF 330 cascaded between the frequency mixer 320 and the frequency doubler 250 is generally disposed outside the chip of the IC. The input frequency of the frequency doubler 250 is equal to the frequency of the output carrier signal delivered through the transmission antenna as shown in FIG. 5. Thus, the carrier signal is fed back through the transmission antenna to the input of the frequency doubler 250 to make the phases of the orthogonal carrier waves unstable, thereby degrading the modulation accuracy.
Those problems of the quadrature modulator of FIG. 3 are more noticeable in the cellular phones which have extremely smaller dimensions.